1. Field of the Invention
The present invention relates to an OFDM receiver and particularly to a method of mode detection for OFDM signals in a DVB-T receiver.
2. Description of the Prior Art
OFDM is a multi-channel modulation system employing Frequency Division Multiplexing (FDM) of orthogonal sub-carriers, each modulated by a low bit-rate digital stream.
In older multi-channel systems using FDM, the total available bandwidth is divided into N non-overlapping frequency sub-channels. Each sub-channel is modulated with a separate symbol stream and the N sub-channels are frequency multiplexed. Even though the prevention of spectral overlapping of sub-carriers reduces (or eliminates) Inter-channel Interference, this leads to an inefficient use of spectrum. The guard bands on either side of each sub-channel waste bandwidth. To overcome the problem of bandwidth wastage, alternatively N overlapping (but orthogonal) sub-carriers, each carrying a baud rate of 1/T and spaced 1/T apart can be used. The sub-carriers are all mathematically orthogonal to each other because of the frequency spacing selected. This permits the proper demodulation of the symbol streams without requiring non-overlapping spectra. Another way of specifying the sub-carrier orthogonality is to require that each sub-carrier have an exact integer number of cycles in the interval T. The modulation of these orthogonal sub-carriers can be represented as an Inverse Fourier Transform. Alternatively, a DFT operation followed by low-pass filtering can generate the OFDM signal. It must be noted that OFDM can be used either as a modulation or multiplexing technique.
The use of Discrete Fourier Transform (DFT) in the parallel transmission of data using Frequency Division Multiplexing was investigated in 1971 by Weinstein and Ebert. In a data sequence d0, d2, . . . , dN−1, where each dn is a complex symbol (the data sequence can be the output of a complex digital modulator, such as QAM, PSK etc), when performing an IDFT on the sequence 2 dn (the factor 2 is used purely for scaling purposes), N complex numbers Sm (m=0,1. . . , N−1) result, as:
                                                                                          S                  m                                =                                ⁢                                  2                  ⁢                                                            ∑                                              n                        =                        0                                                                    N                        -                        1                                                              ⁢                                                                                  ⁢                                                                  d                        n                                            ⁢                                              exp                        ⁡                                                  (                                                      j                            ⁢                                                                                                                  ⁢                            2                            ⁢                            π                            ⁢                                                                                          n                                ⁢                                                                                                                                  ⁢                                m                                                            N                                                                                )                                                                                                                                                                                            =                                ⁢                                  2                  ⁢                                                            ∑                                              n                        =                        0                                                                    N                        -                        1                                                              ⁢                                                                                  ⁢                                                                  d                        n                                            ⁢                                                                        exp                          ⁡                                                      (                                                          j                              ⁢                                                                                                                          ⁢                              2                              ⁢                              π                              ⁢                                                                                                                          ⁢                                                              f                                n                                                            ⁢                                                              t                                m                                                                                      )                                                                          ⁢                                                                                                  [                                                                              m                            =                            0                                                    ,                          1                          ,                                                                                    …                              ⁢                                                                                                                          ⁢                              N                                                        -                            1                                                                          ]                                                                                                                                ⁢                                  ⁢                  where          ,                                    (        2.1        )                                          f          n                =                                            n                              NT                s                                      ⁢                                                  ⁢            and            ⁢                                                  ⁢            t                    =                      mT            s                                              (        2.2        )            
Where, Ts represents the symbol interval of the original symbols. Passing the real part of the symbol sequence represented by equation (2.1) thorough a low-pass filter with each symbol separated by a duration of Ts seconds, yields the signal,
                                          y            ⁡                          (              t              )                                =                      2            ⁢                                                  ⁢            Re            ⁢                          {                                                ∑                                      n                    =                    0                                                        N                    -                    1                                                  ⁢                                                                  ⁢                                                      d                    n                                    ⁢                                      exp                    ⁡                                          (                                              j                        ⁢                                                                                                  ⁢                        2                        ⁢                        π                        ⁢                                                  n                          T                                                ⁢                        t                                            )                                                                                  }                                      ,                              for            ⁢                                                  ⁢            0                    ≦          t          ≦          T                                    (        2.3        )            
Where T is defined as NTs. The signal y(represents the baseband version of the OFDM signal.
It can be noted from (2.3) that the length of the OFDM signal is T, the spacing between the carriers is equal to 1/T, the OFDM symbol-rate is N times the original baud rate, there are N orthogonal sub-carriers in the system, and the signal defined in equation (2.3) is the basic OFDM symbol.
One of the group main advantages of OFDM is its effectiveness against the multi-path delay spread frequently encountered in mobile communication channels. The reduction of the symbol rate by N times results in a proportional reduction of the relative multi-path delay spread, relative to the symbol time. To completely eliminate even the very small ISI that results, a guard time is introduced for each OFDM symbol. The guard time chosen must be larger than the expected delay spread, such that multi-path components from one symbol cannot interfere with the next symbol. Leaving the guard time empty may lead to inter-carrier interference (ICI), since the carriers are no longer orthogonal to each other. To avoid such crosstalk between sub-carriers, the OFDM symbol is cyclically extended in the guard time. This ensures that the delayed replicas of the OFDM symbols always have an integer number of cycles within the FFT interval as long as the multi-path delay spread is less than the guard time.
If the ODFM symbol is generated using equation (2.3), the power spectral density of this signal is similar to that shown in FIG. 1. The sharp-phase transitions caused by phase modulation result in very large side-lobes in the PSD and slow spectrum roll-off (according to a sinc function). If the number of sub-carriers increases, the spectrum roll-off is sharp in the beginning, but degrades further at frequencies further away from the 3-dB cut-off frequency. To overcome the slow spectrum roll-off problem, windowing may be used to reduce the side-lobe level. The most commonly used window is the Raised Cosine Window given by:
      w    ⁡          (      t      )        =      {                                                      0.5              +                              0.5                ⁢                                                                  ⁢                                  cos                  ⁡                                      (                                          π                      +                                              π                        ⁢                                                                                                  ⁢                                                  t                          /                                                      (                                                          β                              ⁢                                                                                                                          ⁢                                                              T                                r                                                                                      )                                                                                                                )                                                                        ,                                                0            ≤            t            ≤                          β              ⁢                                                          ⁢                              T                r                                                                                      1.0            ,                                                              β              ⁢                                                          ⁢                              T                s                                      ≤            t            ≤                          T              r                                                                                                                        0.5                  +                                      0.5                    ⁢                                                                                  ⁢                                          cos                      ⁡                                              (                                                  t                          -                                                      T                            r                                                                          )                                                              ⁢                                          π                      /                      β                                        ⁢                                                                                  ⁢                                          T                      r                                                                      )                            )                        ,                                                              T              s                        ≤            t            ≤                                          (                                  1                  +                  β                                )                            ⁢                              T                r                                                        
Here Tr is the symbol interval chosen to be shorter than the actual OFDM symbol duration, since the symbols are allowed to partially overlap in the roll-off region of the raised cosine window. Incorporating the windowing effect, the OFDM symbol can now be represented as:
            y      ⁡              (        t        )              =          2      ⁢                          ⁢      Re      ⁢              {                              w            ⁡                          (              t              )                                ⁢                                    ∑                              n                =                0                                            N                -                1                                      ⁢                                                  ⁢                                          d                n                            ⁢                              exp                ⁡                                  (                                      j                    ⁢                                                                                  ⁢                    2                    ⁢                    π                    ⁢                                          n                      T                                        ⁢                    t                                    )                                                                    }              ,            for      ⁢                          ⁢      0        ≦    t    ≦    T  
It must be noted that filtering can also be used as a substitute for windowing, for tailoring the spectrum roll-off. Windowing, though, is preferred to filtering because it can be carefully controlled. With filtering, rippling effects in the roll-off region of the OFDM symbol must be avoided. Rippling causes distortions in the OFDM symbol, which directly leads to lower tolerance of display spread?
Based on the previous discussion, the method for generating an ODFM symbol is as follows.
First, the N input complex symbols are padded with zeros to get Ns symbols to calculate the IFFT. The output of the IFFT is the basic OFDM symbol.
Based on the delay spread of the multi-path channel, a specific guard-time must be chosen (e.g. Tg). A number of samples corresponding to this guard time must be taken from the beginning of the OFDM symbol and appended to the end of the symbol. Likewise, the same number of samples must be taken from the end of the OFDM symbol and inserted at the beginning.
The OFDM symbol must be multiplied by the raised cosine window to remove the power of the out-of-band sub-carriers.
The windowed OFDM symbol is then added to the output of the previous OFDM symbol with a delay of Tr, so that there is an overlapping region of βTr between each symbol.
OFDM system design, as with any other system design, involves tradeoffs and conflicting requirements. The following are the most important design parameters of an OFDM system and may form part of a general OFDM system specification. Bit Rate required for the system, available bandwidth, BER requirements (power efficiency) and RMS delay spread of the channel.
Guard Time
Guard time in an OFDM system usually results in an SNR loss in an OFDM system, as it carries no information. The choice of the guard time is straightforward once the multi-path delay spread is known. In general practice, the guard time must be at least 2-4 times the RMS delay spread of the multi-path channel. Further, higher-order modulation schemes (like 32 or 64 QAM) are more sensitive to ISI and ICI than simple schemes like QPSK. This factor must also be taken into account when determining the guard-time.
Symbol Duration
To minimize SNR loss due to guard time, symbol duration must be set much higher than guard time. An increase in symbol time, however, implies a corresponding increase in the number of sub-carriers and thus an increase in the system complexity. A practical design choice for symbol time requires at least five times the guard time, which yields acceptable SNR loss.
Number of Sub-carriers
Once the symbol duration is determined, the number of sub-carriers required can be determined by first calculating the sub-carrier spacing by simply inverting the symbol time (less the guard period). The number of sub-carriers is the available bandwidth divided by the sub-carrier spacing.
Modulation and Coding Choices
The first step in selecting coding and modulation techniques is to determine the number of bits carried by an OFDM symbol. A suitable combination of modulation and coding techniques can then be selected to fit the input data rate into the OFDM symbols and, simultaneously satisfy the bit-error rate requirements. Selection of modulation and coding techniques is simplified, as each channel is assumed to be almost AWGN and there is no requirement for consideration of the effects of multi-path delay spread.
OFDM possesses inherent advantages for wireless communications.
As discussed earlier, the increase in the symbol time of the OFDM symbol by N times (N being the number of sub-carriers), leads to a corresponding increase in the effectiveness of OFDM against the ISI caused due to multi-path delay spread. Further, use of the cyclic extension process and proper design can completely eliminate ISI from the system.
In addition to delay variations in the channel, the lack of amplitude flatness in the frequency response of the channel also causes ISI in digital communication systems. A typical example would be use of twisted-pair cable in telephone lines. These transmission lines are designed for voice calls and offer poor frequency response in high frequency transmission. In systems that use single-carrier transmission, an equalizer may be required to mitigate the effect of channel distortion. The complexity of the equalizer depends upon the severity of the channel distortion and there are frequently issues such as equalizer non-linearities and error propagation etc. that present additional obstacles.
In OFDM systems, on the other hand, since the bandwidth occupied by each sub-carrier is very small, the amplitude response over such narrow bandwidth will be basically flat (it can be safely assumed that the phase response will be linear over said narrow bandwidth). In the case of extreme amplitude distortion, an equalizer of very simple structure will be sufficient to correct distortion in each sub-carrier.
The use of sub-carrier modulation improves the flexibility of OFDM to combat channel fading and distortion, and makes it possible for the system to transmit at maximum possible capacity using the technique of channel loading. If the transmission channel has a fading notch in a certain frequency range corresponding to a certain sub-carrier, the presence of this notch can be detected using channel estimation schemes, and assuming that the notch does not vary rapidly enough compared to the symbol duration of the OFDM symbol, it is possible to change (scale down/up) the modulation and coding schemes for this particular sub-carrier (i.e., increase their robustness against noise), so that overall sub-carrier capacity is maximized. This however requires data from channel-estimation algorithms. In the case of single-carrier systems, no remedy exists for preventing such fading notches. The data must somehow survive the distortion using error correction coding or equalizers.
Impulse noise usually comprises a burst of interference in channels such as the return path HFC (Hybrid-Fiber-Coaxial), twisted-pair and wireless channels caused by lightning or other atmospheric phenomena. It is common for the length of the interference waveform to exceed the symbol duration of a typical digital communication system. For example, in a 10 MBPS system, the symbol duration is 0.1 μs, and an impulse noise waveform, of about two micro-seconds, can cause a burst of errors that cannot be corrected using normal error-correction coding. Usually complicated Reed-Solomon codes in conjunction with huge interleaves are used to correct this problem. OFDM systems are inherently robust against impulse noise, since the symbol duration of an OFDM signal is much longer than that of the corresponding single-carrier system and thus, it is less likely that impulse noise will cause (even single) symbol errors. Thus, complicated error-control coding and interleaving schemes for handling burst-type errors are not required for OFDM systems, thus simplifying transceiver design.
OFDM is the best environment in which to employ frequency diversity. In fact, in a combination of OFDM and CDMA, called MC-CDMA transmission, frequency diversity is inherently present in the system (i.e., it is freely available). Although OFDM provides advantages for wireless transmission, it also has some serious disadvantages that must be overcome for this technology to succeed.
Many applications that use OFDM technology have arisen in the last few years. In the following, one such application is described in detail.
Digital Video Broadcasting (DVB) is a standard for broadcasting Digital Television over satellite, cable, and terrestrial (wireless) transmission.
Approved by the DVB Steering Board in December 1995, DVB-T is the system specification for the terrestrial broadcast of digital television signals. This work was based on a set of user requirements produced by the Terrestrial Commercial Module of the DVB project. DVB members contributed to the technical development of DVB-T through the DTTV-SA (Digital Terrestrial Television-Systems Aspects) of the Technical Module. The European Projects SPECTRE, STERNE, HD-DIVINE, HDTVT, dTTb, and several other organizations developed system hardware and produced test results that were fed back to DTTV-SA.
As with the other DVB standards, MPEG-2 audio and video encoding forms the transmitted data of DVB-T. Other elements of the specification include a transmission scheme based on orthogonal frequency-division multiplexing (OFDM), which allows for the use of either 1705 carriers (usually known as 2 k), or 6817 carriers (8 k). Concatenated error correction is used. The 2 k mode is suitable for single-transmitter operation and for relatively small single-frequency networks with limited transmitter power. The 8 k mode can be used both for single-transmitter operation and for large-area single-frequency networks. The guard interval is selectable. As well, Reed-Solomon outer coding and outer convolutional interleaving are used, as with the other DVB standards, and another error-correction system, using a punctured convolutional code, is added. The amount of overhead required by the inner code of the second error-correction system, can be adjusted to suit the needs of the service provider. The data carriers in the coded orthogonal frequency-division multiplexing (COFDM) frame can use QPSK and different levels of QAM modulation and code rates to trade bits for enhanced robustness. Bi-level hierarchical channel coding and modulation can be used, but hierarchical source coding is not used. The latter was deemed unnecessary by the DVB group because its benefits did not justify the extra receiver complexity. Finally, the modulation system combines OFDM with QPSK/QAM. OFDM uses a large number of carriers that spread the information content of the signal. Used successfully in DAB (digital audio broadcasting), the major advantage of OFDM is resistance to multi-path interference.
Improved immunity to multi-path interference is obtained through the use of a guard interval, a portion of the digital signal given away for echo resistance. This guard interval reduces the transmission capacity of OFDM systems. However, the greater the number of OFDM carriers provided, for a given maximum echo time delay, the less transmission capacity is lost. Nonetheless, a tradeoff is involved. Simply increasing the number of carriers has a significantly detrimental impact on receiver complexity and phase-noise sensitivity.
Because of the multi-path immunity of OFDM, it may be possible to operate an overlapping network of transmitting stations with a single frequency. In the areas of overlap, the weaker of the two received signals is similar to an echo signal. However, if the two transmitters are far apart, causing a large time delay between the two signals, the system will require a large guard interval.
The potential exists for three different operating environments for digital terrestrial television in Europe, including broadcast on a currently unused channel, such as an adjacent channel, or on a clear channel; broadcast in a small-area single-frequency network (SFN); or broadcast in a large-area SFN.
One of the group main challenges for the DVB-T developers is that the different operating environments lead to somewhat different optimum OFDM systems. The common 2 k/8 k specification has been developed to offer solutions for all (or nearly all) operating environments.
It should be noted that, in the DVB-T system, the ratio of guard interval Tg over the desired symbol interval Tu may be 1/32, 1/16, ⅛and ¼, and Tu is respectively 2048 and 8192 in the 2K-mode and 8K-mode transmission. Thus, in order to recover the original information carried in an OFDM signal received from an OFDM transmitter, the values of Tu and Tg must be known before implementing guard interval removal and discrete Fourier transformation. A mode detection mechanism is thus required in the DVB-T receiver.
In U.S. Pat. No. 6,330,293, Otto Klank et al. disclose a mode detection method. At the receiver end, coarse time synchronization linked to mode detection and, possibly and additionally, coarse AFC (automatic frequency correction) are carried out initially both for seeking and identifying received signals, as well as for continuously monitoring them. The time signal is correlated with the time signal shifted by the desired symbol length Tu. This correlation may be carried out more than once, for example five times per data frame. In this correlation, signal samples of different length Tu are used, depending on the respective mode, and the maximum correlation result obtained from this are then used to deduce the present mode (for example 2K or 8K modes). If no usable correlation result maximum is obtained, the correlation steps may be repeated.
FIG. 2 is a diagram showing a mode detector disclosed in U.S. patent application publication No. 2002/0186791. The I and Q samples of the received signal are supplied to an input terminal 10. The samples are supplied to a 2 k and 8 k size first-in first-out (FIFO) memory 121 and 122. The moving average correlation of the samples over a minimum guard period is then calculated in blocks 141 and 142, and the power of the correlation measured in blocks 161 and 162. The correlation function is calculated in blocks 141 and 142 by multiplying input symbols with symbols contemporaneously obtained from the delay blocks 121 and 122 with the delay applied thereto, thereby obtaining a measure of the correlation between them. The results are then summed, and a running average is calculated over a number of samples, equal to the smallest allowed guard interval size, that is, 1/32 of the FFT size. Thus, for example, g=64 and 256 samples in 2 k and 8 k mode respectively. Each combination of the blocks 141 and 161, and 142 and 162 therefore forms a correlation function, and the separation between peaks in each correlation function depends on the total duration of the symbol plus the guard period. The resulting measurements are passed to blocks 181 and 182 for decimation (i.e., removal of some portion of the samples). The samples remaining after decimation in blocks 181 and 182 are then passed through filtering resonators 191-198, each centered at a respective resonance frequency based on the COFDM symbol frequency of a particular combination of the mode and the guard interval. A counter (not shown) is provided at the output of each of the resonators 191-198, and each counter increments when its peak power is largest. The peak power produced by each resonator is then compared. Thus, by examining the counter values after a number of symbols, the counter with the highest value is determined to be that which corresponds to the mode (either 2 k or 8 k) and guard period used by the transmitted signal.
However, the mode detection using only correlation result maxima or power peak is susceptible to noise. Multi-path propagation reduces the correlation result maxima or power peak, and makes it indistinct. Thus, no usable correlation result maxima or power peak will be obtained or detected if the RF signal is received through multi-path propagation. Further, the correlation operation requires a large memory to temporarily store the symbols of the OFDM signal.